Motors whose each phase coil is wound around stator poles in full pitch and distributed winding have been widely known. FIG. 34 is a partially axial cross section schematically illustrating an example of the structure of such a conventional motor.
In FIG. 34, the conventional motor is provided with a substantially annular shaped stator core 4, a motor housing 6 in which the stator core 4 is installed, and a pair of coil end portions 5 of stator windings installed in the stator core 4. The motor is also provided with a substantially annular shaped rotor core 2 rotatably disposed inside the stator core 4 with a gap therebetween, and a rotor shaft 1 fixed to the inner periphery of the rotor core 2 and rotatably supported by the motor housing 6 with a pair of bearings 3.
FIG. 37 is a cross sectional view taken on line AA-AA in FIG. 34. In these FIGS. 34 and 37, a four-pole, 24-slot synchronous reluctance motor is illustrated. A full pitch and distributed winding is used as each of the three-phase stator winding. Each of U-, V-, and W-phase coils is distributedly wound around corresponding stator poles at 180-degree pitches in electric angle. In FIG. 37, reference character 35J represents a back yoke of the stator, and reference character 35H represents teeth of the stator.
In one area of 360 degrees electric angle of the stator between the first slot {circle around (1)} and the twelfth slot {circle around (12)}, reference characters 351 and 352 represent a +U-phase winding, and reference characters 357 and 358 represent a −U-phase winding. The +U-phase winding and −U-phase winding are installed in the corresponding slots {circle around (1)}, {circle around (2)}, {circle around (7)}, and {circle around (8)} of the stator to form a first U-phase coil. In the specification, these signs “+” and “−” represents a reversed phase therebetween.
Similarly, referenced characters 355 and 356 represent a +V-phase winding, and reference characters 35B and 35C represent a −V-phase winding. The +V-phase winding and −V-phase winding are installed in the corresponding slots {circle around (5)}, {circle around (6)}, {circle around (11)}, and {circle around (12)} of the stator to form a first V-phase coil.
In addition, referenced characters 359 and 35A represent a +W-phase winding, and reference characters 353 and 354 represent a −W-phase winding. The +W-phase winding and −W-phase winding are installed in the corresponding slots {circle around (9)}, {circle around (10)}, {circle around (3)}, and {circle around (4)} of the stator to form a first W-phase coil.
As well as the one area of 360 degrees electric angle of the stator, in the other area of 360 degrees electric angle of the stator between the thirteenth slot and the twenty-fourth slot, a second U-phase coil, a second V-phase coil, and a second W-phase coil are formed.
Each of the rotor core 2 and the stator core 4 is composed of a plurality of magnetic steed sheets laminated in an axial direction of the rotor shaft 1.
The rotor core 2 has a salient structure. Specifically, the rotor core 2 is formed with first to fourth groups of chordal flux barriers 35F punched out in slit by press working. The first to fourth groups of the flux barriers 35F are symmetrically arranged with respect to the axial direction of the rotor shaft 1 such that:
each of the first to fourth groups of the flux barriers 35F is circumferentially spaced apart from another adjacent group thereof;
the flux barriers of each of the first to fourth groups are aligned in a corresponding radial direction of the rotor core 2 at intervals therebetween; and
both ends of each of the flux barriers of each of the first to fourth groups extend toward the outer periphery of the rotor core 2 with predetermined thin edges thereof left between the both ends and the outer periphery.
The first to fourth groups of the flux barriers 35F provide thin magnetic paths 35E therebetween. The thin edges of the rotor core 2 are continued to each other, this supports the thin magnetic paths 35E.
A direct axis (d-axis) and a quadrature axis (q-axis) are normally defined in the rotor as a rotating coordinate system (rotor coordinate system); these d-q axes are rotated as the rotor is rotated. The d-axis has a high magnetic permeability, and the q-axis has a low magnetic permeability because of the flux barriers 35F.
The configuration of the motor illustrated in FIG. 37 creates a reluctance torque based on the difference between the magnetic impedance in the d-axis and that in the q-axis, thus rotating the rotor (rotor core 2 and the rotor shaft 1).
FIG. 39 is a block diagram schematically illustrating an example of the circuit structure of a control system for relatively precisely controlling such a motor.
The control system illustrated in FIG. 39 includes an encoder (E) 592, an interface (E-IF) 593, a current sensor (not shown), and a converter 59H.
The encoder 592 detects information indicative of a rotational position θr and a rotational speed (angular velocity) ω of a motor (rotor) 591. The interface (E-IF) 593 for the encoder 592 converts the detected information to the rotational position θr and rotational speed ω of the motor 591 and passing them to the converter 59H.
The current sensor detects instantaneous U- and W-phase winding currents iu and iw respectively flowing through the U-phase winding and W-phase winding of the stator of the motor 591.
The converter 59H converts a stationary coordinate system (u-v-w coordinate system) into the d-q coordinate system. Specifically, the converter 59H receives the instantaneous U- and W-phase winding currents iu and iw passed from the current sensor and the rotational position θr of the rotor passed from the interface 593 and converts the instantaneous U- and W-phase winding currents iu and iw into instantaneous d- and q-axis current components id and iq on respective d- and q-axis of the d-q coordinate system based on the rotational position θr of the rotor.
The control system includes a speed difference detector 594, a speed controller 595, and a command current determiner 596.
The speed difference detector 594 receives a command indicative of a target speed ω* of the motor 591 and externally input thereto, and subtracts the detected rotational speed ω from the target speed ω* to detect a speed difference therebetween.
The speed controller 595 receives the speed difference detected by the speed error detector 594, and executes a compensating operation by calculating a proportional term and an integral term based on the calculated speed difference so as to obtain a torque demand T*. The command current determiner 596 determines a d-axis command current id* and q-axis command current iq* based on the torque demand T* and the detected rotational speed ω.
The control system includes a feedforward voltage determiner 597 and a current-control loop gain determiner 59B.
The voltage signal generator 597 receives the determined d-axis command current id*, q-axis command current iq*, and the detected rotational speed ω, and determines a d-axis feedforward voltage command FFd and q-axis feedforward voltage command FFq based on the received d-axis command current id* q-axis command current iq*, and the detected rotational speed ω.
The current-control loop gain determiner 59B has stored therein a loop gain Gd for a d-axis current control loop and a loop gain Gq for a q-axis current control loop; these loop gains Gd and loop gain Gq have determined by default.
The control system includes a d-axis current difference detector 598, a d-axis current controller 599, and a d-axis voltage controller 59A.
The d-axis current difference detector 598 subtracts the d-axis current component id from the d-axis command current iq* to calculate a d-axis current difference therebetween.
The d-axis current controller 599 receives the d-axis current difference calculated by the d-axis current difference detector 598. The d-axis current controller 599 executes a compensating operation by calculating a proportional term and an integral term based on the received d-axis current difference so as to obtain a d-axis current control voltage command proportional to the current-loop gain Gd. The obtained d-axis current control voltage command is passed to the d-axis voltage controller 59A.
Similarly, the control system includes a q-axis current difference detector 59C, a q-axis current controller 59D, and a q-axis voltage controller 59E.
The q-axis current difference detector 59C subtracts the q-axis current component iq from the q-axis command current iq* to calculate a q-axis current difference therebetween.
The q-axis current controller 59D receives the q-axis current difference calculated by the q-axis current difference detector 59C. The q-axis current controller 59D executes a compensating operation by calculating a proportional term and an integral term based on the received q-axis current difference so as to obtain a q-axis current control voltage command proportional to the current-loop gain Gq. The obtained q-axis current control voltage command is passed to the q-axis voltage controller 59E.
The control system includes a converter 59F and a three-phase inverter 59G.
The d-axis voltage controller 59A serves as an adder. Specifically, the d-axis voltage controller 59A calculates the sum of the d-axis current control voltage command passed from the d-axis current controller 599 and the d-axis feedforward voltage command FFd. In addition, the d-axis voltage controller 59A passes, as a d-axis command voltage vd*, the calculated sum of the d-axis current control voltage command and the d-axis feedforward voltage command FFd to the converter 59F.
Similarly, the q-axis voltage controller 59E serves as an adder. Specifically, the q-axis voltage controller 59E calculates the sum of the q-axis current control voltage command passed from the q-axis current controller 59D and the q-axis feedforward voltage command FFq. In addition, the q-axis voltage controller 59E passes, as a q-axis command voltage vq*, the calculated sum of the q-axis current control voltage command and the q-axis feedforward voltage command FFq to the converter 59F.
The converter 59F converts the d-axis command voltage vd* and q-axis command voltage vq* on the respective d and q awes into U-, V-, and W-phase voltage commands vu*, vv*, and vw* in the stationary coordinate system, and outputs the converted U-, V-, and W-phase voltage commands vu*, vv*, and vw* to the three-phase inverter 59G.
FIG. 38 is a circuit diagram schematically illustrating an example of the structure of the three-phase inverter 59G.
The three-phase inverter 59G is composed of a direct current (DC) battery N95, a first pair of series-connected power semiconductor elements N96 and N9A, a second pair of series-connected power semiconductor elements N97 and N9B, and a third pair of series-connected power semiconductor elements N98 and N9C. As the power semiconductor elements, power transistors, such as IGBTs (Insulated Gate Bipolar Transistors) or MOSFETs can be preferably used, respectively.
For example, the first pair (N96 and N9A), second pair (N97 and N9B), and third pair (N98 and N9C) of power semiconductor elements are parallely connected to each other in bridge configuration.
A connecting point through which the power semiconductor elements of each pair are connected to each other in series is connected to an output lead extending from the other end of a corresponding one of the U-, V-, and W-phase winding of the motor 591.
One end of the series-connected power semiconductor elements of each pair is connected to a positive terminal of the DC battery N95, and the other end thereof is connected to a negative terminal thereof.
Each of the power transistor elements N96, N97, N98, N9A, N9B, and N9C is individually driven ON and OFF based on a corresponding PWM (Pulse Width Modulation) drive signal input thereto. This allows a higher DC voltage of the DC battery N95 to be chopped so that U-, V-, and W-phase voltages corresponding to the U-, V-, and W-phase voltage commands vu*, vv*, and vw* are supplied to the U-, V-, and W-phase windings N92, N93, and N94 of the motor 591, respectively (see FIG. 38).
The duty cycles of the PWM drive signals to be supplied to the respective power transistor elements N96, N97, N98, N9A, N9B, and N9C are individually controlled. This can control the U-, V-, and W-phase voltages to be supplied to the U-, V-, and W-phase windings N92, N93, and N94 of the motor 591 to thereby control the rotational speed and the output of the motor 591.
Note that the functional blocks illustrated in FIG. 39 except for the motor 591, the three-phase inverter 59G, the encoder 592, and the interface 593 can be implemented by tasks to be executable by a microprocessor (microcomputer) in accordance with a program.
Such a control system, for example, illustrated in FIG. 39, is disclosed in U.S. Pat. No. 6,954,050 corresponding to Japanese Patent Application Publication No. 2004-289959. As described above, a microprocessor-based control system designed to execute the motor output control based on the rotating coordinate system (d-q coordinate system) has been normally used as motor control systems.
Control for various types of motors including such a synchronous reluctance motor and an interior permanent magnet motor is normally executed by such a control system illustrated in FIG. 39.
However, characteristic curves of a motor with respect to armature current are not represented as ideally linear curves. This may result various problems associated with the motor output.
Such various problems associated with the nonlinear motor characteristic curves will be described hereinafter.
As described above, the rotor core and the stator core of a motor are normally comprised of a plurality of soft magnetic steel sheets laminated in their thickness directions. The rotor core is formed with a plurality of permanent magnets installed therein as needed. One of causes that make the motor control difficult is a nonlinear magnetic property of the soft magnetic steel sheets.
As is generally known, a magnetic steel sheet has a nonlinear magnetic property; this nonlinear magnetic property represents that a magnetic steel sheet has a magnetic saturation property. In order to reduce a motor in size and in manufacturing cost, an armature current range in which a nonlinear characteristic curve of the motor is saturated due to the magnetic saturation property of the magnetic steel sheets is used to control the motor output.
Specifically, such control of the motor output is executed based on control parameters, such as a d-axis inductance Ld and a q-axis inductance Lq, required to execute the motor output control on the supposition that the control parameters are each constant despite of the nonlinear characteristic curves of the motor. This may cause the motor output to be inaccurately controlled when the armature current lies within a nonlinear region of a characteristic curve of the motor; there is the possibility that control errors occur in the motor-output control routines.
For example, in the block diagram of FIG. 39, when the armature current lies within a nonlinear region of a magnetic property curve of the motor, the control system cannot address:
the control sensitivity of the d-axis current controller 599 is changed depending on the magnitude of the armature current;
the control sensitivity of the q-axis current controller 59D is changed depending on the magnitude of the armature current; and
the d- and q-axis feedforward voltage commands FFd and FFq each represent an inaccurate value.
In addition, the control system also cannot meet that the d-axis command current id* and q-axis command current iq* determined by the command current determiner 596 each represent an inaccurate value due to the nonlinearity of the magnetic property of the output torque of the motor.
When the armature current lies within a specific linear portion of a characteristic curve of the motor, such as a motor speed curve or motor output torque curve thereof, therefore, the motor output can be properly controlled. However, when the armature current lies within a nonlinear region of the characteristic curve of the motor, the motor output can be improperly controlled.
Specifically, it may be difficult for the control system to properly control, in the whole region of the motor speed curve or of the motor-output torque curve, the three-phase voltages outputted from the inverter 59G and supplied to the three-phase windings of the motor.
In order to address the problems set forth above, the conventional control system illustrated in FIG. 39 is designed to compensate the three-phase voltages based on the fed-back actual rotational speed of the motor, the fed-back instantaneous d-axis current id, and the fed-back q-axis current iq by eliminating:
the difference between the fed-back actual rotational speed of the motor and the target speed;
the difference between the fed-back d-axis current component id and the d-axis command current id*; and
the difference between the fed-back q-axis current component iq and the q-axis command current iq*.
However, when high-speed responsibility is required for the motor control and/or a motor is controlled to be rapidly rotated, the range of the three-phase voltages output from the inverter 59G that can be compensated by the feedback control set forth above is limited. This may not meet the need of the high-speed responsibility of the motor and also may not meet the rapid rotation of the motor.
When permanent magnets are additionally installed in a rotor core of a motor, a magnetic flux created by each of the permanent magnets depends on magnetomotive force based on the armature current and applied to the rotor core. This may cause the motor control to become complicated.
In recent years, in order to reduce such a motor control system in manufacturing cost and improve the reliability thereof, various types of sensorless (encoder-less) motor control have been widely used. In the various types of sensorless motor control, however, because no encoder (rotational position sensor) is provided therein, it may be difficult to properly grasp the rotational position of the rotor.